I approached this project as a complete engineering exercise: from the operating principles of a photovoltaic module to the final grid-connected test of a 1 kW single-phase solar inverter. My goal was to design a solar inverter that extracts as much energy as possible from a PV array, converts that energy through a boost stage and a full-bridge inverter, and delivers a sinusoidal current into the utility grid with correct frequency and phase. Throughout the design, the solar inverter is treated not as an isolated power stage but as the central interface between a nonlinear, environment-dependent source and a stiff grid. The work therefore combines power electronics, analog signal conditioning, digital control, maximum power point tracking, and synchronization.
Motivation and Context
Energy and environmental pressures have made renewable generation a central topic in modern power engineering. Among renewable sources, solar energy is attractive because it is widely distributed, clean at the point of use, and available in small as well as large installations. A grid-connected photovoltaic system is often the preferred architecture because it avoids the cost, volume, and maintenance burden of battery banks. In such a system, the solar inverter is the key power electronic converter. Its efficiency, reliability, and control quality determine how much of the available solar energy actually reaches the grid.
I studied the main solar inverter topologies before choosing a structure for this design. The main candidates are the direct inverter, the line-frequency isolated inverter, the high-frequency isolated inverter, and the high-frequency non-isolated boost inverter. Each has advantages and disadvantages. The direct topology is simple and efficient but has no galvanic isolation. The line-frequency isolated topology provides isolation and robustness but is heavy and costly. The high-frequency isolated topology reduces transformer size but introduces additional switching and thermal challenges. The high-frequency non-isolated boost topology offers a good compromise for low-power single-phase systems: it is compact, relatively efficient, and allows the PV input voltage to vary over a wide range.
| Topology | Isolation | Efficiency | Weight | Typical Application |
|---|---|---|---|---|
| Direct inverter | No | Very high | Low | Transformerless grid-tied systems |
| Line-frequency isolated | Yes | Moderate | High | Industrial and larger systems |
| High-frequency isolated | Yes | Moderate to high | Medium | Low-power residential systems |
| High-frequency non-isolated boost | No | High | Low | Single-phase residential solar inverter |
For this project, I selected the high-frequency non-isolated boost solar inverter architecture. It uses a front-end DC-DC boost converter and a rear-end DC-AC full-bridge inverter. The boost stage raises the PV voltage to a stable DC-link voltage, while the inverter stage produces a sinusoidal AC current. This two-stage arrangement gives me independent control of maximum power point tracking and grid current shaping. The solar inverter therefore has a clear division of labor: the boost stage manages the source, and the inverter stage manages the grid.
Solar Cell Characteristics
The behavior of a solar inverter cannot be understood without the behavior of its source. A photovoltaic cell is essentially a large p-n junction. When photons with sufficient energy reach the junction, electron-hole pairs are generated, and an electric field separates the charges. The result is a current source whose magnitude depends on irradiance and temperature. The single-diode model is widely used to describe this behavior:
$$I = I_L – I_0 \left[ \exp\left( \frac{q(V + I R_s)}{n k T} \right) – 1 \right] – \frac{V + I R_s}{R_{sh}}$$
In this expression, \(I_L\) is the photocurrent, \(I_0\) is the diode saturation current, \(q\) is the electron charge, \(V\) is the terminal voltage, \(I\) is the terminal current, \(R_s\) is the series resistance, \(R_{sh}\) is the shunt resistance, \(n\) is the ideality factor, \(k\) is Boltzmann’s constant, and \(T\) is the absolute temperature. The thermal voltage is
$$V_T = \frac{kT}{q}$$
The saturation current and photocurrent vary with temperature and irradiance. A useful approximation for the photocurrent is
$$I_L(G,T) = \frac{G}{G_{ref}} I_{sc,ref} \left[ 1 + \alpha (T – T_{ref}) \right]$$
where \(G\) is irradiance, \(G_{ref}\) is the reference irradiance, \(I_{sc,ref}\) is the short-circuit current at reference conditions, and \(\alpha\) is the temperature coefficient of short-circuit current. The saturation current can be modeled as
$$I_0(T) = I_0(T_{ref}) \left( \frac{T}{T_{ref}} \right)^3 \exp\left[ \frac{q E_g}{n k} \left( \frac{1}{T_{ref}} – \frac{1}{T} \right) \right]$$
where \(E_g\) is the bandgap energy. These equations explain why the maximum power point of a PV module moves when irradiance or temperature changes. For a typical module, the short-circuit current is almost proportional to irradiance, while the open-circuit voltage falls as temperature rises. The maximum power point voltage is usually about 80% of the open-circuit voltage under standard conditions.
| Parameter | Symbol | Typical Value in This Design |
|---|---|---|
| Open-circuit voltage | \(V_{oc}\) | 21.0 V |
| Short-circuit current | \(I_{sc}\) | 3.74 A |
| Maximum power voltage | \(V_{mp}\) | 17.1 V |
| Maximum power current | \(I_{mp}\) | 3.8 A |
| Maximum power | \(P_{mp}\) | 59.9 W |
I used these characteristics to build a simulation model of the PV source. The model showed that the PV output is a current-source-like region at low voltage and a voltage-source-like region near open circuit. Between these regions, there is a single maximum power point for a given irradiance and temperature. The solar inverter must continuously locate this point, because the available power changes with weather and cell temperature.
Maximum Power Point Tracking Principle
Maximum power point tracking is one of the most important functions of a solar inverter. It is also one of the most visible ways in which the solar inverter affects the economics of a PV system. If the operating voltage is too low or too high, the array produces less power than it could. The control problem is to adjust the converter so that the PV source sees an equivalent load resistance that matches its internal resistance at the maximum power point.
For a boost converter, the ideal voltage relationship is
$$\frac{V_{out}}{V_{in}} = \frac{1}{1-D}$$
where \(D\) is the duty ratio. If the output load resistance is \(R_L\), the equivalent input resistance seen by the PV source is approximately
$$R’_{L} = R_L (1-D)^2$$
By changing \(D\), I can change the equivalent input resistance. When \(R’_L\) matches the source resistance at the maximum power point, the array transfers maximum power. This is the basic impedance-matching idea behind maximum power point tracking. In practice, the solar inverter measures PV voltage and current, computes power, and adjusts the duty ratio so that power moves toward its peak.
Overall System Architecture
The solar inverter I designed is organized into several functional units: the auxiliary power supply, the main control unit, the boost and inverter power stage, the gate driver unit, and the sampling unit. The auxiliary supply provides the low-voltage rails required by the controller, drivers, and signal-conditioning circuits. The main control unit is built around a TMS320F2812 digital signal controller. The power stage contains a front-end boost converter and a rear-end full-bridge inverter. The driver unit converts low-power PWM signals into gate signals for the IGBTs. The sampling unit measures PV voltage and current, inverter output current, and grid voltage.
The system operates as follows. After power-up, the controller initializes its peripherals, timers, ADC, GPIO, and interrupt vectors. The PV voltage and current are sampled. The boost converter is then controlled so that the PV array operates near its maximum power point, and the DC-link voltage is regulated to about 400 V. The full-bridge inverter converts this DC-link voltage into a sinusoidal AC current. The output current passes through an LCL filter, and the resulting current is synchronized with the grid voltage. The sampling unit continuously feeds the controller with measurements, and the controller adjusts the PWM signals to maintain stable operation.
| Unit | Main Function | Key Components |
|---|---|---|
| Auxiliary power | Provide +3.3 V, +5 V, +15 V, -15 V | Transformer, rectifiers, 7815, 7915, 7805, 1117-3.3 |
| Main control | Execute MPPT, SPWM, PLL, protection | TMS320F2812, reset, crystal, JTAG |
| Boost stage | Raise PV voltage to DC link | IGBT, inductor, diode, DC-link capacitor |
| Inverter stage | Convert DC link to AC current | Four IGBTs, anti-parallel diodes |
| Driver unit | Amplify and isolate gate signals | IR2110, bootstrap capacitor, bootstrap diode |
| Sampling unit | Measure voltages and currents | Hall sensors, op-amp conditioning |
Hardware Design
I designed the hardware in a modular way so that each part of the solar inverter could be tested and debugged independently. The power stage, control stage, and signal-conditioning stage are separated as much as possible to reduce noise coupling. The auxiliary supply is also separated because it contains a line-frequency transformer and rectifiers that can inject ripple into sensitive analog circuits.
Auxiliary Power Supply
The auxiliary power supply converts 220 V AC into several isolated DC rails. A transformer steps the voltage down to about 17 V AC. A rectifier and filter produce an unregulated DC bus. Positive and negative linear regulators generate +15 V and -15 V for operational amplifiers and Hall sensors. A +5 V regulator supplies the gate driver logic and other digital circuits. Finally, a low-dropout regulator produces +3.3 V for the DSP core and I/O. The +3.3 V rail is especially important because it powers the main controller. Any instability on this rail would directly affect the PWM outputs and the ADC measurements.
| Rail | Load | Regulator | Filtering |
|---|---|---|---|
| +15 V | Op-amps, Hall sensors | 7815 | Electrolytic and ceramic capacitors |
| -15 V | Op-amps, Hall sensors | 7915 | Electrolytic and ceramic capacitors |
| +5 V | IR2110 logic, digital interfaces | 7805 | Electrolytic and ceramic capacitors |
| +3.3 V | DSP2812 core and I/O | 1117-3.3 | Ceramic and tantalum capacitors |
Front-End Boost Converter
The front-end boost converter raises the PV voltage to the DC-link level. I used an IGBT as the main switch, a fast recovery diode as the boost diode, an inductor for energy storage, and a large capacitor at the output. The inductor current ramps up when the IGBT is on and ramps down when the IGBT is off. The diode prevents the output capacitor from discharging back into the switch. The output capacitor supplies the inverter stage during the switch-on interval. This is the classic boost action, and it allows the solar inverter to accept a wide PV input voltage range.
The inductor value is chosen from the allowed current ripple. A simplified design equation is
$$L = \frac{V_{PV} D T_s}{\Delta I_L}$$
where \(V_{PV}\) is the PV voltage, \(D\) is the duty ratio, \(T_s\) is the switching period, and \(\Delta I_L\) is the peak-to-peak inductor current ripple. The ripple coefficient is
$$\eta = \frac{\Delta I_L}{I_L}$$
For this design, I used a switching frequency of 20 kHz and allowed a current ripple of about 25%. The calculated inductance was approximately 2.548 mH, and I selected 3.0 mH to provide margin. The larger value reduces peak current stress and improves the quality of the current waveform, though it increases the physical size of the inductor.
The output capacitor of the boost stage must hold the DC link while the inverter draws pulsating current. Its value can be estimated from the allowable voltage ripple:
$$C_{dc} = \frac{P_o \alpha T_s}{\eta U_{dc}^2}$$
where \(P_o\) is the output power, \(\alpha\) is the duty ratio, \(T_s\) is the switching period, \(\eta\) is the voltage ripple coefficient, and \(U_{dc}\) is the DC-link voltage. Using \(P_o = 1000\) W, \(U_{dc} = 400\) V, \(T_s = 50 \mu s\), \(\alpha = 0.7\), and a ripple coefficient of 2%, the calculated capacitance was about 875 \(\mu F\). I selected a 4000 \(\mu F\), 450 V capacitor to provide a large safety margin and to reduce DC-link voltage fluctuations during load changes.
| Boost Component | Design Value | Selected Value | Reason |
|---|---|---|---|
| Inductor | 2.548 mH | 3.0 mH | Lower ripple and margin |
| Output capacitor | 875 \(\mu F\) | 4000 \(\mu F\) | DC-link stability |
| Switching frequency | 20 kHz | 20 kHz | Filter size and loss balance |
| DC-link voltage | 400 V | 400 V | Inverter modulation requirement |
Full-Bridge Inverter Stage
The rear-end inverter is a single-phase full bridge. It contains four IGBTs with anti-parallel fast recovery diodes. The DC link from the boost stage feeds the bridge. By switching the diagonal pairs of the bridge in a sinusoidal PWM pattern, the inverter produces a high-frequency pulse train whose fundamental component is a 50 Hz sine wave. The output filter removes the switching harmonics. The anti-parallel diodes provide a path for inductive current when the switches turn off, protecting the IGBTs and allowing reactive current flow.
I used a unipolar or hybrid switching scheme in which one leg switches at high frequency while the other leg switches at line frequency. This reduces switching losses because two of the four IGBTs do not switch at 20 kHz. It also reduces the harmonic content compared with a simple bipolar scheme. The important point is that the solar inverter must never short the DC link. Therefore, the gate signals include dead time, and the control logic prevents complementary conduction in the same leg.
LCL Output Filter
The output filter is critical for grid current quality. A pure L filter is simple, but it requires a large inductance to attenuate switching harmonics. An LC filter is useful in stand-alone mode but can interact with the grid in grid-connected mode. An LCL filter provides better attenuation with smaller components. I selected an LCL filter for the solar inverter because it reduces the switching-frequency current ripple while keeping the total inductance reasonable.
For an LCL filter, the transfer function from inverter voltage to grid current can be written in simplified form as
$$\frac{I_d(s)}{U_d(s)} = \frac{1}{s(L_1 + L_2) + s^3 L_1 L_2 C_f}$$
where \(L_1\) is the inverter-side inductor, \(L_2\) is the grid-side inductor, and \(C_f\) is the filter capacitor. The filter resonates at
$$f_{res} = \frac{1}{2\pi} \sqrt{\frac{L_1 + L_2}{L_1 L_2 C_f}}$$
I placed the resonant frequency between the line frequency and the switching frequency. For a 20 kHz switching frequency and a 50 Hz line frequency, this gives enough separation. In my design, I used a total inductance of about 4 mH and a capacitor in the range of a few microfarads. The exact values were chosen after simulation and experimental adjustment. The LCL filter makes the output current smooth and reduces the high-frequency content that would otherwise be injected into the grid.
| Filter Parameter | Symbol | Design Consideration |
|---|---|---|
| Inverter-side inductance | \(L_1\) | Limits current ripple |
| Grid-side inductance | \(L_2\) | Reduces grid-side harmonics |
| Filter capacitance | \(C_f\) | Sets resonant frequency |
| Resonant frequency | \(f_{res}\) | Between line and switching frequency |
Gate Driver Design
The gate driver must turn the IGBTs on and off reliably while protecting the controller from high voltages and currents. I used the IR2110 high-voltage high-speed driver. It includes level shifting, under-voltage lockout, and a bootstrap circuit for the high-side switch. The bootstrap capacitor supplies the high-side gate charge, and the bootstrap diode prevents reverse current. The driver also provides isolation between the low-power control ground and the power stage. This is important because the solar inverter has fast switching edges and high common-mode noise.
Each IR2110 can drive two switches in one bridge leg. Because the full bridge has four IGBTs, I used two IR2110 devices. The boost converter requires an additional driver channel. The gate resistors were chosen to control the switching speed. A resistor that is too small causes fast edges, ringing, and electromagnetic interference. A resistor that is too large increases switching loss and reduces efficiency. I selected values that gave a good compromise between switching loss and noise.
Main Control Unit
The main control unit is based on the TMS320F2812. I chose this device because it has a fast 32-bit CPU, a hardware multiplier, event managers for PWM generation, a 12-bit ADC, capture units for grid synchronization, and enough GPIO for protection and status signals. The solar inverter needs all of these features. The event manager generates the PWM signals for the boost converter and the full-bridge inverter. The ADC samples the PV voltage, PV current, output current, and grid voltage. The capture unit detects zero crossings of the grid voltage for phase locking.
The controller board includes a reset circuit, a crystal oscillator, a JTAG interface, and decoupling capacitors. The reset circuit holds the DSP in a known state during power-up. The crystal provides the system clock. The JTAG interface allows program download and real-time debugging. I paid special attention to the analog power supply and ground layout because the ADC accuracy depends on a clean reference and low noise. The digital and analog grounds are connected at a single point to avoid ground loops.
| Controller Feature | Use in Solar Inverter |
|---|---|
| Event manager PWM | Boost gate signal and full-bridge SPWM |
| 12-bit ADC | PV voltage, PV current, output current, grid voltage |
| Capture unit | Grid zero-crossing detection |
| CPU timer | Control loop timing and MPPT interval |
| GPIO | Protection, relay control, status indication |
Sampling Circuits
The sampling circuits convert high-voltage and high-current signals into low-voltage signals that the DSP ADC can read. I used Hall-effect sensors because they provide isolation and good accuracy. The PV voltage sensor measures the array voltage. The PV current sensor measures the array current. The output current sensor measures the AC current injected into the grid. The grid voltage sensor measures the AC line voltage for synchronization and protection.
Each sensor output is conditioned by an operational amplifier circuit. The conditioning circuit performs scaling, offset shifting, and low-pass filtering. The DSP ADC input range is 0 V to 3.3 V, so the signals must be biased around 1.65 V for bipolar AC measurements. I used a voltage divider and a precision reference to create this bias. The low-pass filter removes high-frequency noise that could alias into the ADC result. I also added clamping diodes to protect the ADC pins from overvoltage.
| Signal | Sensor Type | Purpose | ADC Channel |
|---|---|---|---|
| PV voltage | Hall voltage sensor | MPPT and protection | ADCINA0 |
| PV current | Hall current sensor | MPPT and protection | ADCINA1 |
| Output current | Hall current sensor | Current control and protection | ADCINA2 |
| Grid voltage | Hall voltage sensor | PLL and protection | ADCINA3 |
Software Design
I developed the software in a modular way. The main program initializes the system and then enters a background loop. The control work is done in interrupt service routines. The most important interrupts are the PWM period interrupt, the ADC interrupt, and the capture interrupt. The PWM period interrupt updates the sinusoidal modulation index and the boost duty ratio. The ADC interrupt reads the sampled values and performs filtering and scaling. The capture interrupt detects the grid zero crossing and resets the sine table index so that the output current remains synchronized with the grid.
The main program performs the following steps:
| Step | Action |
|---|---|
| 1 | Initialize system control, PLL, watchdog, and clocks |
| 2 | Disable interrupts and clear interrupt flags |
| 3 | Initialize PIE control and vector table |
| 4 | Configure GPIO for PWM, ADC, and capture |
| 5 | Initialize ADC, event manager, and timers |
| 6 | Enable interrupts and start timers |
| 7 | Enter background loop for monitoring and protection |
SPWM Generation
I used regular sampling to generate the sinusoidal PWM for the solar inverter. In regular sampling, the sine wave is sampled at fixed intervals, and the sampled value determines the pulse width for that switching period. This method is simpler than natural sampling and is well suited to a digital controller. The on-time for the \(k\)-th switching period is
$$t_{on} = \frac{T_c}{4} \left[ 1 + M \sin\left( \frac{2\pi k}{N} \right) \right]$$
where \(T_c\) is the carrier period, \(M\) is the modulation index, \(N\) is the number of samples per fundamental cycle, and \(k = 0, 1, \ldots, N-1\). The sine table is stored in memory, and the PWM compare register is updated every switching period. By changing the table index, I can adjust the output frequency and phase. By changing \(M\), I can adjust the output current amplitude.
The DSP event manager generates the PWM signals with dead time. Dead time is necessary to prevent shoot-through in each bridge leg. I set the dead time according to the IGBT turn-off time and the driver delay. If the dead time is too short, the bridge can short the DC link. If it is too long, the output waveform distortion increases. The correct value was found experimentally by observing the gate signals and the output current.
Software Phase-Locked Loop
The solar inverter must inject current in phase with the grid voltage. I implemented a software phase-locked loop using the capture unit and the ADC. The grid voltage is conditioned into a square wave. The capture unit detects the rising edge, which corresponds to the positive zero crossing. The controller then resets the sine table index. The ADC also samples the grid voltage, and the controller compares the phase of the output current with the phase of the grid voltage. If the output current leads the grid voltage, the table index is advanced more slowly. If the output current lags, the table index is advanced more quickly. This adjusts the phase until the error is small.
The PLL algorithm also checks the grid frequency. If the measured frequency is outside the allowed range, the solar inverter disconnects from the grid. This protects the inverter and prevents islanding. The capture interrupt is generated by the zero-crossing signal. A simple digital filter rejects spurious edges caused by noise. The PLL therefore provides both synchronization and protection.
| PLL Function | Implementation |
|---|---|
| Zero-crossing detection | Capture unit on rising edge |
| Frequency measurement | Timer counts between captures |
| Phase correction | Adjust sine table index |
| Islanding protection | Disconnect if frequency or voltage is out of range |
Maximum Power Point Tracking Algorithm
I studied several maximum power point tracking methods before choosing a perturb-and-observe approach. The perturb-and-observe method is popular because it is simple, requires only voltage and current measurements, and does not need a detailed PV model. However, the conventional method has two weaknesses. First, the perturbation step is fixed, so there is a trade-off between tracking speed and steady-state ripple. Second, the operating point continues to oscillate around the maximum power point, which causes small power losses and current fluctuations.
In the conventional perturb-and-observe method, the controller measures PV voltage and current, computes power, and compares the new power with the previous power:
$$\Delta P = P(k) – P(k-1)$$
$$P(k) = V(k) I(k)$$
$$\Delta V = V(k) – V(k-1)$$
If \(\Delta P > 0\), the duty ratio is changed in the same direction as the previous step. If \(\Delta P < 0\), the duty ratio is changed in the opposite direction. This continues until the operating point reaches the maximum power point. The flow is simple, but the fixed step means that a small step gives slow tracking and a large step gives large oscillations.
First Improvement: Variable Step Size
My first improvement was to make the perturbation step proportional to the slope of the power-voltage curve. On the left side of the maximum power point, \(dP/dV > 0\). On the right side, \(dP/dV < 0\). The absolute value of the slope is large far from the maximum power point and small near it. Therefore, I used the ratio \(\Delta P / \Delta V\) to set the step size:
$$\Delta U_{ref} = K \frac{\Delta P}{\Delta V}$$
I limited the step size to avoid instability. The left-side limit was set larger than the right-side limit because the power-voltage curve is flatter on the left for the module I modeled. In my design, the maximum left-side step was 2, and the maximum right-side step was -1. This variable-step method reaches the maximum power point faster than the fixed-step method and reduces steady-state oscillation.
| Algorithm | Step Size | Tracking Time | Steady-State Ripple |
|---|---|---|---|
| Conventional perturb-and-observe | Fixed | About 0.10 ms in simulation | Relatively high |
| First improvement: variable step | Proportional to slope | About 0.04 ms in simulation | Lower |
| Second improvement: intermittent tracking | Variable with pause | Similar to variable step | Very low |
| Third improvement: combined initial voltage | Variable with initialized operating point | About 0.01 ms in simulation | Very low |
Second Improvement: Intermittent Tracking
My second improvement was to stop perturbing once the maximum power point had been reached, as long as the environmental conditions remained stable. I monitored the change in power. If the power change remained below a small threshold, the controller kept the current duty ratio constant. If the power change exceeded the threshold, the controller restarted the tracking process. This intermittent method reduces steady-state ripple and avoids unnecessary perturbations. It also reduces the computational load on the solar inverter controller.
The threshold must be chosen carefully. If it is too small, noise can trigger unnecessary tracking. If it is too large, the controller may miss a real change in irradiance or temperature. I used a threshold based on the measured power and the ADC resolution. In simulation, the power remained within about 0.05 W after the maximum power point was reached. This was small enough to be considered stable.
Third Improvement: Initial Voltage from Constant-Voltage Estimation
My third improvement addressed the startup behavior. In a conventional perturb-and-observe system, the initial operating voltage is often zero or far from the maximum power point. This means that the controller spends time searching from a poor starting point. I combined the perturb-and-observe method with a constant-voltage initialization. The constant-voltage method assumes that the maximum power point voltage is a fixed fraction of the open-circuit voltage, typically about 80%. However, the open-circuit voltage changes with temperature. Therefore, I used a temperature-dependent estimate of the maximum power point voltage.
At startup, the controller sets the initial reference voltage to the estimated maximum power point voltage instead of zero. This places the operating point near the maximum power point immediately. Then the variable-step perturb-and-observe method fine-tunes the duty ratio. The combination reduces startup time and improves the dynamic response. In simulation, the tracking time was reduced to about 0.01 ms, which is much faster than the conventional method. The improvement is particularly useful when the solar inverter starts in the morning, when irradiance is changing quickly.
Interrupt and Control Flow
The control software uses three main interrupt sources. The PWM period interrupt updates the PWM compare values. The ADC interrupt reads the sampled signals and performs the control calculations. The capture interrupt handles grid synchronization. The ADC interrupt is triggered by the event manager at the end of each conversion. Inside the ADC interrupt, I perform scaling, offset correction, and digital filtering. I then run the MPPT algorithm at a slower rate than the current control loop. This is important because the PV source has a slow thermal and electrical time constant compared with the grid cycle.
The current control loop uses the sampled output current and the grid voltage phase. It calculates the error between the desired current and the actual current and adjusts the modulation index. The desired current amplitude is derived from the DC-link voltage controller, which balances the power from the PV side and the power delivered to the grid. If the DC-link voltage rises, the current amplitude reference increases. If the DC-link voltage falls, the current amplitude reference decreases. This keeps the DC-link voltage within a safe range.
| Interrupt | Trigger | Main Tasks |
|---|---|---|
| PWM period interrupt | Timer 1 underflow | Update sine table index and boost duty |
| ADC interrupt | End of ADC conversion | Read samples, filter, run MPPT and current control |
| Capture interrupt | Grid zero crossing | Reset table index, measure frequency, PLL correction |
Protection Functions
A practical solar inverter must include protection. I implemented overvoltage and undervoltage protection for the PV input, overcurrent protection for the boost inductor and inverter output, DC-link overvoltage protection, and grid abnormal protection. The DSP compares the sampled values with software thresholds. If a fault is detected, the PWM outputs are disabled, and the grid relay is opened if necessary. I also included a watchdog timer to reset the controller if the software hangs. The protection logic is executed in the background loop and in the ADC interrupt, so the response is fast enough for the power stage.
| Protection | Measured Signal | Action |
|---|---|---|
| PV overvoltage | PV voltage | Disable boost PWM |
| PV undervoltage | PV voltage | Reduce power or shut down |
| Boost overcurrent | PV current | Disable boost PWM |
| Inverter overcurrent | Output current | Disable inverter PWM, open relay |
| DC-link overvoltage | DC-link voltage | Disable inverter PWM |
| Grid abnormal | Grid voltage and frequency | Open relay, stop current injection |
System Performance Testing
After completing the hardware and software, I built a prototype and tested it in several stages. The first test was the auxiliary power supply. The second test was the boost PWM. The third test was the SPWM waveform. The fourth test was the inverter output current. The final test was grid connection. Each test was performed with oscilloscopes, multimeters, and current probes. The purpose was to verify that the solar inverter met its design targets and that the control loops were stable.
Auxiliary Power Supply Test
The auxiliary power supply was tested before connecting the main power stage. The +3.3 V rail was measured first because it powers the DSP. The +5 V rail was tested next because it supplies the gate driver logic. The +15 V and -15 V rails were tested last because they supply the analog signal conditioning circuits. The measured values were close to the design targets. The +3.3 V rail had no measurable deviation. The +5 V rail had a small deviation of about 0.01 V. The +15 V rail had a deviation of about 0.34 V, which is acceptable for the operational amplifiers and Hall sensors. These results showed that the auxiliary supply was stable enough for the solar inverter to operate.
| Rail | Target | Measured | Deviation | Result |
|---|---|---|---|---|
| +3.3 V | 3.30 V | 3.30 V | 0.00 V | Pass |
| +5 V | 5.00 V | 4.99 V | 0.01 V | Pass |
| +15 V | 15.00 V | 14.66 V | 0.34 V | Pass |
| -15 V | -15.00 V | -14.70 V | 0.30 V | Pass |
Boost PWM Test
The boost PWM signal was measured at the gate driver input. The frequency was 20.0017 kHz, which is very close to the designed 20 kHz. The small frequency error is caused by the crystal tolerance and the timer resolution. The duty ratio was changing during the test because the maximum power point tracking algorithm was active. The duty ratio increased gradually, which indicated that the controller was adjusting the operating voltage in the correct direction. This confirmed that the boost control loop was working and that the solar inverter was responding to the PV source.
SPWM Waveform Test
The SPWM waveform was measured at the output of the event manager. The frequency was 20.0012 kHz, again very close to the design value. The pulse widths followed a sinusoidal pattern, which is the expected behavior for regular sampling. The dead time was visible between the complementary signals. The waveform was clean, with no abnormal jitter. This test confirmed that the inverter modulation was correct and that the sine table was being read properly.
| Signal | Designed Frequency | Measured Frequency | Deviation |
|---|---|---|---|
| Boost PWM | 20 kHz | 20.0017 kHz | 1.7 Hz |
| Inverter SPWM | 20 kHz | 20.0012 kHz | 1.2 Hz |
Inverter Output Current Test
The inverter output current was measured with a current probe and an oscilloscope. The waveform was a sine wave at 49.9994 Hz. The small frequency deviation is acceptable for grid connection. The waveform showed a small amount of distortion, which is caused by the dead time, the switching harmonics, and the non-ideal behavior of the filter. However, the total harmonic distortion was within an acceptable range for the prototype. The current amplitude responded correctly when the load or the DC-link voltage changed. This confirmed that the current control loop was functioning.
Grid-Connected Test
The final test connected the solar inverter to the grid through an isolation transformer and a protection relay. I measured both the inverter output current and the grid voltage simultaneously. The phase difference between the current and the voltage was very small, which means that the solar inverter was injecting current at nearly unity power factor. The frequency of the output current matched the grid frequency. The PLL maintained synchronization even when the grid voltage changed slightly. The solar inverter remained stable during the test, and the protection system responded correctly when the grid voltage was intentionally disturbed.
| Test Quantity | Result | Acceptance Criterion |
|---|---|---|
| Output current frequency | 49.9994 Hz | 50 Hz \(\pm\) 0.5 Hz |
| Output current phase | Nearly in phase with grid voltage | Power factor close to unity |
| Grid voltage synchronization | Stable | No runaway or loss of lock |
| DC-link voltage | About 400 V | Stable within design range |
| Protection response | Correct | PWM disabled on fault |
Discussion of Results
The test results show that the solar inverter design is fundamentally sound. The auxiliary power supply provides stable rails. The boost converter raises the PV voltage to the required DC-link level. The full-bridge inverter produces a sinusoidal current. The LCL filter reduces switching harmonics. The DSP controller executes the MPPT algorithm, the SPWM generation, and the PLL in real time. The maximum power point tracking improvements reduce tracking time and steady-state ripple. The grid-connected test confirms that the output current is synchronized with the grid voltage.
There are still opportunities for improvement. The efficiency of the solar inverter could be increased by using silicon carbide or gallium nitride devices. The filter could be optimized to reduce size and cost. The MPPT algorithm could be made more robust under partial shading by combining it with a global search method. The control loop could include repetitive control or proportional-resonant control to reduce harmonic distortion further. The communication interface could be added for remote monitoring and fault diagnosis. These improvements would make the solar inverter more suitable for commercial deployment.
Conclusion
I designed, built, and tested a 1 kW single-phase grid-connected solar inverter. The solar inverter uses a two-stage architecture with a boost converter and a full-bridge inverter. The boost stage performs maximum power point tracking and regulates the DC link. The inverter stage produces a sinusoidal current and synchronizes with the grid. The controller is based on the TMS320F2812, and the software includes ADC sampling, SPWM generation, PLL, protection, and an improved perturb-and-observe MPPT algorithm.
The improved MPPT algorithm uses variable step size, intermittent tracking, and a temperature-dependent initial voltage estimate. Simulation results show that the tracking time is reduced from about 0.10 ms to about 0.01 ms compared with the conventional method. The steady-state ripple is also reduced. The hardware tests show that the auxiliary power supply, boost PWM, SPWM, output current, and grid synchronization all meet the design requirements. The solar inverter therefore provides a practical foundation for further research and development in single-phase photovoltaic grid-connected power conversion.
In summary, the solar inverter is the critical component that connects a photovoltaic source to the grid. Its design requires careful attention to power topology, component selection, control algorithms, and protection. The work presented here demonstrates that a digital-controlled two-stage solar inverter can achieve stable maximum power point tracking and high-quality grid current injection. The experience gained from this design can be applied to larger solar inverter systems, multi-string architectures, and advanced grid-support functions.